1. Field of the Invention
The present invention relates generally to the field of digital wireless communications. Specifically, the present invention relates to architectures of a digitally controlled radio frequency (RF) transmitter system. More specifically, the present invention relates to transmitter architectures that can operate in different modulation standards and different frequency bands by using substantially same building blocks.
2. Description of the Related Art
The fast growing demand for mobile communication systems has created many modulation (or speed) standards involving various rates and ways devices communicate with each other. These common modulation standards, such as GSM and the third generation (3G) for mobile radio standards, have compelled transmitter designers to combine different modulation standards in one device while maintaining high linearity, high power efficiency and low cost.
Particularly, the power efficiency is of great importance to portable devices (e.g., cellular phones and palm computers) employing different modulation standards, since the power amplifier is a main determinant of battery talk time. The requirements of high linearity, high power efficiency and low cost result in a strong restriction on the implementation of a RF transmitter which must have both ultra-linearity and high power efficiency for constant and non-constant envelope modulation standards.
Conventional RF transmitter architectures follow a linear approach in which the modulation of a power amplifier is performed in a linear region. However, such an operation leads to a significant decrease in the power efficiency. In order to achieve both high linearity and high power efficiency, the power efficient non-linear switching-mode power amplifier has been used with linearization techniques.
There are at least three different linearization approaches: the Linear Amplification with Nonlinear Components (LINCs) technique, the Envelope Elimination and Restoration (EER) technique and predistortion. The basic principle of LINCs technique, as illustrated in FIG. 1 and described in the article “Linear Amplification using Nonlinear Components” by D. C. Cox, IEEE Trans. on Communications, Vol. COM-22, pages 1942 to 1945, December 1974, is to take an envelope modulated bandpass signal and decompose it through a signal decomposer 102 into two out-phased constant envelop signals that are respectively applied to a pair of highly nonlinear power efficient amplifiers 104 (PA1) and 106 (PA2). The outputs from the amplifiers 104 and 106 are summed through a passive combiner 108. The major advantage of the LINC transmitter 100 is that the RF amplification is performed by two highly efficient non-linear amplifiers (e.g., class-C, D or E), each operating on constant envelope signals. However, one of the disadvantages of this technique is the extremely tight tolerances on the matching of the two amplifier paths to achieve acceptable small out-of-band rejection. Another one of the drawbacks encountered with this technique is that the output power combiner will introduce a significant loss of power efficiency.
The Envelop Elimination and Restoration (EER) technique, as illustrated in FIG. 2 and described in the article “Single-sideband transmission by envelope elimination and restoration” by Leonard. R. Kahn, Proceedings of the IRE, Vol. 40, pages 803 to 806, July 1952, is to apply an input RF signal to an Envelope Detector 210 to generate an amplitude-modulated signal, and a Hard Limiter 212 to generate a phase-modulated signal. Since the RF power amplifier 216 is freed from the requirement to amplify non-constant envelope signals, the amplitude-modulated signal from the Envelope Detector 210 can control the amplitude of the switching-mode power amplifier 216 through an adjustment of its bias and supply voltage via an amplitude controller 214. As the phase-modulated signal from Hard Limiter 212 has a constant envelope, it can thus be implemented by one of the switching-mode RF power amplifier classes (e.g. Class D, E or F) that feature very high DC to RF power conversion efficiency. However, any mismatch between the analog phase and the amplitude signals must be minimized. This is difficult since the two paths include vastly different types of circuit elements, working at widely different frequencies.
Predistortion is a well-known concept where the input signal is modified in order to compensate for the distortion introduced by a power amplifier when it is operating in nonlinear mode. The basic form of a predistortion linearization scheme is shown in FIG. 3. The predistorter 320 with a function P(Vin) operates on the input signal Vin in such a manner that its output signal X is distorted in a precisely complementary manner to the distortion produced by the power amplifier 322 with a transfer function F(X). As a result, the output signal Vout is therefore, ideally, an amplified, but undistorted replica of the input signal with a transfer function F(P(Vin)). The task of the adaptive predistortion is to adaptively predistort the input samples in order to minimize the errors. The switching-mode power amplifier has inherent non-linearities associated with them. These non-linearities are due to the fact that the amplifier's gain and phase characteristics change with a change in the power supply voltage, and thus the signal dependent power supply voltage is a source of amplitude to amplitude (AM-AM) and amplitude to phase (AM-PM) distortion. The adaptive predistortion adjusts the modulation signal by inverse characteristic of the power amplifier. Thus, the overall response of the cascaded predistorter and power amplifier demonstrates linearity. Some experimental results and demonstrating are given in the article “Experimental Performance of an Adaptive Digital Linearized Power Amplifier” by Andrew S. Wright and Willem G. Durtler, IEEE Transactions on Vehicular Technology, Vol. 41, No. 4, pages 395-400, November 1992. The main advantage of the digital predistortion is its capability for wide bandwidth operation and simplicity for the implementation.
One example of direct phase modulation is to use a fractional-N frequency synthesizer 400 as illustrated in FIG. 4. But the bandwith of the fractional-N frequency synthesizer 400 is strongly limited by its phase-looked loop formed by a divider 404, a phase detector 403, a loop filter 402 and a VCO 401. In particular, the article “A 27-mW CMOS Fractional-N Synthesizer Using Digital Compensation for 2.5-Mb/s GFSK Modulation” by Michael H. Perrott and etc., IEEE Journal of Solid-State Circuits, Vol. 32, No. 12, December 1997, pages 2048 to 2060, describes a wideband fractional-N synthesizer. It requires a compensated transmit filter which is difficult to implement, and an accurate loop filter to mitigate the mismatch problem. However, this approach is not efficient and suitable for the phase modulation with different modulation standards.
There is thus a great need for a generic RF transmitter architecture that can maximize the power efficiency with wide bandwidth and high linearity for different modulation standards at low cost.